反相放大器 The Inverting Amplifier

The basic operational amplifier circuit is shown in Figure 1. This circuit gives closed-loop gain of R2/R1 when this ratio is small compared with the amplifier open-loop gain and, as the name implies, is an inverting circuit. The input impedance is equal to R1. The closed-loop bandwidth is equal to the unity-gain frequency divided by one plus the closed-loop gain.
The only cautions to be observed are that R3 should be chosen to be equal to the parallel combination of R1 and R2 to minimize the offset voltage error due to bias current and that there will be an offset voltage at the amplifier output equal to closed-loop gain times the offset voltage at the amplifier input.
Offset voltage at the input of an operational amplifier is comprised of two components, these components are identified in specifying the amplifier as input offset voltage and input bias current. The input offset voltage is fixed for a particular amplifier, however the contribution due to input bias current is dependent on the circuit configuration used. For minimum offset voltage at the amplifier input without circuit adjustment the source resistance for both inputs should be equal. In this case the maximum offset voltage would be the algebraic sum of amplifier offset voltage and the voltage drop across the source resistance due to offset current. Amplifier offset voltage is the predominant error term for low source resistances and offset current causes the main error for high source resistances.
In high source resistance applications, offset voltage at the amplifier output may be adjusted by adjusting the value of R3 and using the variation in voltage drop across it as an input offset voltage trim.
Offset voltage at the amplifier output is not as important in AC coupled applications. Here the only consideration is that any offset voltage at the output reduces the peak to peak linear output swing of the amplifier.
The gain-frequency characteristic of the amplifier and its feedback network must be such that oscillation does not occur. To meet this condition, the phase shift through amplifier and feedback network must never exceed 180° for any frequency where the gain of the amplifier and its feedback network is greater than unity. In practical applications, the phase shift should not approach 180° since this is the situation of conditional stability. Obviously the most critical case occurs when the attenuation of the feedback network is zero. Amplifiers

which are not internally compensated may be used to achieve increased performance in circuits where feedback network attenuation is high. As an example, the LM101 may be operated at unity gain in the inverting amplifier circuit with a 15 pF compensating capacitor, since the feedback network has an attenuation of 6 dB, while it requires 30 pF in the non-inverting unity gain connection where the feedback network has zero attenuation. Since amplifier slew rate is dependent on compensation, the LM101 slew rate in the inverting unity gain connection will be twice that for the non-inverting connection and the inverting gain of ten connection will yield eleven times the slew rate of the non-inverting unity gain connection. The compensation trade-off for a particular connection is stability versus bandwidth, larger values of compensation capacitor yield greater stability and lower bandwidth and vice versa.
The preceding discussion of offset voltage, bias current and stability is applicable to most amplifier applications and will be referenced in later sections. A more complete treatment is contained in Reference 4.

基本的反相放大器電路示于圖1。顧名思義，在R2/R1的數(shù)值遠(yuǎn)小于OP開環(huán)增益時，這個數(shù)值就是反相放大器的增益。輸入阻抗就是R1的阻值，閉環(huán)增益帶寬=單位增益帶寬/（1+閉環(huán)增益）
在設(shè)計中需要注意的是：R3的阻值應(yīng)該等于R1和R2的并聯(lián)電阻，以減小輸入偏置電流所帶來的失調(diào)電壓。輸出失調(diào)電壓=輸入失調(diào)電壓*閉環(huán)增益
OP輸入端的失調(diào)電壓有兩個主要的來源，輸入偏置電流（Input bias current）和輸入失調(diào)電壓（Input offset voltage）。對于一個給定的OP，輸入失調(diào)電壓就已經(jīng)確定了，但是由輸入失調(diào)電流所帶來的失調(diào)電壓與所采用的電路結(jié)構(gòu)有關(guān)系，為了在不使用調(diào)整電路的情況下減小輸入偏置電流所帶來的失調(diào)電壓，應(yīng)該使得同、反相輸入端對地的直流電阻相等，使得由于偏置電流在輸入電阻上壓降所帶來的失調(diào)電壓相互抵消。在對低內(nèi)阻信號源的放大器中，OP的輸入失調(diào)電壓將成為失調(diào)電壓誤差的主要來源，而對于高內(nèi)阻信號源的放大器，OP的輸入偏置電流在信號源內(nèi)阻上的壓降將成為上述誤差的主要來源。
在高輸入阻抗的情況下，失調(diào)電壓可以采用R3的阻值來調(diào)整，利用輸入偏置電流在其上的壓降來對輸入失調(diào)電壓作出補償（既用這個的得到的壓降來抵消輸入失調(diào)電壓）。
在交流耦合的時候，失調(diào)電壓并不顯得很重要，這時的主要問題是：失調(diào)電壓減小了輸出電壓峰—峰值（P-P）的線性動態(tài)范圍。
工作在閉環(huán)狀態(tài)下的OP和其反饋網(wǎng)絡(luò)的增益——頻率特性，必須保證不會產(chǎn)生震蕩。為了實現(xiàn)穩(wěn)定，OP和反饋環(huán)路對任何頻率的信號，在環(huán)路增益大于1時的環(huán)路相移角度絕對不能超過180°。在實踐上，為了達(dá)到穩(wěn)定條件，相移角度不應(yīng)該接近180°。顯然，最為臨界的狀態(tài)發(fā)生在反饋網(wǎng)絡(luò)的衰減為0的時候（既跟隨器狀態(tài)）。沒有做內(nèi)部補償?shù)腛P可以被用在深度反饋電路中以取得更好的頻率特性。舉一個例子：LM101在反相單位增益電路中，由于反饋網(wǎng)絡(luò)6dB的衰減，僅需要15pF的補償電容；但在同相單位增益電路中，由于反饋環(huán)路沒有衰減，就需要30pF的補償電容。由于OP的轉(zhuǎn)換速度（SR）取決于補償電容的大小， LM101的SR在反相單位增益電路里可以做到同相單位增益電路的2倍。反相電路在同等SR的條件下可以做到跟隨器電路的110%。對于一個給定的OP放大器電路，在進(jìn)行補償時需要在穩(wěn)定性和帶寬之間進(jìn)行進(jìn)行權(quán)衡。加大補償電容可以提高穩(wěn)定性，但是犧牲了放大器的增益帶寬，反之亦然。